The high-precision measurement of an object's temperature is an important, technically subtle problem, which in the prior art required expensive, difficult engineering tradeoffs. Principal applications include sensing a point versus imaging a line or area, and contact sensing versus remote sensing. An object 's temperature can be measured directly by a sensor in thermally conductive contact with it, convectively by a sensor measuring a fluid in thermally conductive contact with it, and/or radiatively whereby the object's black body radiation is measured by a sensor that does not require direct physical contact and may be remote.
Reference is now made to FIG. 1A, which shows a prior-art temperature measurement device known as a resistance temperature detector or resistance temperature device (RTD). The resistance of the RTD is a function of temperature, so the temperature of the RTD can be determined by simply measuring its resistance, and the change in temperature of the RTD can be determined by measuring the change in resistance. RTDs can use any number of materials which exhibit a change in resistance as a function of temperature, including metals and semiconductors. Platinum is commonly used, with commercial devices made to have a standard resistance at a fixed temperature such as 100 Ω when the RTD temperature is 0° C., and a change in resistance of 0.385 Ω per kelvin, resulting in a resistance coefficient of 0.385% per kelvin. Such RTD sensors promise an accuracy of 0.13 kelvins when measuring a temperature near 0° C., and generally exhibit degraded performance for higher and lower temperatures, unless additional calibrations are made.
In FIG. 1A, the RTD element 1 is connected to a Kelvin bridge circuit. Lead wire 2 connects to one side of the RTD element 1 and lead wire 3 connects to the other side of the RTD element 1 as shown in the figure. Lead wires 2 and 3 should be kept as short as practical, as resistance in these wires may affect the measurement of the RTD temperature. Wires 2A and 2B are connected to wire 2 and wires 3A and 3B are connected to wire 3. Wires 2A and 3A are used to force a current through RTD element 1, with the current being provided by current source 4. Wires 2B and 3B are used to sense the voltage induced on lead wires 2 and 3. The voltage induced on lead wires 2 and 3 is caused by the current flow through the series connection of wire 2, RTD element 1, and wire 3, and follows Ohm's Law, where the voltage is the product of the total resistance and the current produced by current source 4. In a properly designed Kelvin bridge circuit, the current flow through wires 2B and 3B is negligible, such that the voltage drop along wires 2B and 3B is a negligible fraction of the voltage dropped across the RTD element 1, enabling an accurate measurement of the RTD resistance to be made even when wires 2A, 2B, 3A, and 3B are long. Those skilled in the art will recognize that the 4-wire Kelvin bridge can be replaced by a 2-wire resistance measurement provided that the error induced by the resistance of the lead wires is negligible compared to the RTD resistance and the desired accuracy.
Reference is now made to FIG. 1B, which show another prior-art temperature sensor using a silicon semiconductor diode instead of a RTD. The forward-biased voltage across a diode biased to a constant current has a temperature coefficient of about 2.3 mV per kelvin and is reasonably linear. Without calibration, the measurement accuracy of a silicon diode sensor can be as large as ±30 kelvins, since the measured voltage depends on details of the diode area, structure, and non-ideality. Calibrated diodes can achieve an accuracy approaching ±1 kelvins or better. If absolute temperature measurements without calibration are a requirement, one can use two identical silicon diodes, usually monolithically integrated, and operate the diodes by forcing currents I1 and I2 through diodes D1 and D2 respectively, and measuring voltages V1 and V2 across diodes D1 and D2 respectively. The absolute temperature can then be calculated from equation 1 below to achieve an accuracy of about 1 kelvin or better without calibration.T=(V1−V2)/(8.7248×10−5 ln(I1/I2))   (equation 1)
In FIG. 1B, diode D1 is represented by symbol 11A, diode D2 is represented by symbol 11B, current I1 is represented by symbol 14A, current I2 is represented by symbol 14B, voltage V1 is measured by voltmeter 15A, and voltage V2 is measured by voltmeter 15B. Lead wires 12A and 13A are connected directly to diode D1, and lead wires 12B and 13B are connected directly to diode D2. To minimize the resistive voltage drops across lead wires 12A and 13A, a 4-wire Kelvin connection of wires 16A, 16B, 17A, and 17B is used to connect to diode D1. Similarly, to minimize the resistive voltage drops across lead wires 12B and 13B, a 4-wire Kelvin connection of wires 18A, 18B, 18A, and 18B is used to connect to diode D1. Wires 16A and 17A are connected to current source I1. Wires 16B and 17B are connected to the voltmeter 15A. Wires 18A and 19A are connected to current source I2. Wires 18B and 19B are connected to voltmeter 15B. Those skilled in the art will recognize that the diode measurements can force current and measure voltage, as described above. Alternatively, the diode measurement can force voltage and measure current. Note that these prior art measurements of temperature rely on changes in the forward-bias current of diodes, which can be estimated using the ideal diode equation and the temperature dependence of the forward voltage or forward current on the intrinsic carrier concentration.
Remote temperature measurements and power sensors generally fall into a class of detectors called bolometers and calorimeters. A bolometer typically measures the power of incident electromagnetic signal by converting the irradiance into thermal power, and subsequently measuring a corresponding change in temperature. Calorimeters typically measure the energy of incident electromagnetic signals, usually in pulsed measurements and in special cases with a resolution that allows the measurement of the energy of a single photon. Unlike photonic detectors, bolometers measure power absorbed, and hence can be designed to work with any wavelength of incident electromagnetic signals, including (but not limited to): radio wave, microwave, mm wave, THz waves, infrared (including but not limited to) far infrared, long wavelength infrared (LWIR), mid wavelength infrared (MWIR), short wavelength infrared (SWIR), near infrared (NIR)), visible, ultraviolet, X-ray, and gamma ray. Bolometers use a variety of techniques to absorb the electromagnetic energy, including direct absorption in a material; antenna-coupled absorbers (where the temperature rise of a load impedance on the antenna is measured); and a small-area absorber illuminated by an optical system (e.g. lenses & refractors (notably including glass and GRIN lenses), reflectors, gratings & Fresnel lenses, etc.) to achieve a higher temperature rise by concentrating the incident energy on a smaller mass. The term microbolometer refers to a small-area bolometer (typically used in arrays for imaging applications) formed using fabrication techniques from the microelectronics industry (such as lithography, additive processes like deposition, and subtractive processes like etching) that are often collectively termed microelectromechanical systems or MEMs. Numerous microbolometers have been described empirically and theoretically in the prior art. Some significant examples include the following, included herein by reference:
Uncooled Microbolometers:
For example, see                Paul W. Kruse, Uncooled Thermal Imaging Arrays, Systems, and Applications (SPIE Tutorial Texts in Optical Engineering TT51, 2001).        Paul W. Kruse, “Uncooled IR Focal Plane Arrays,” Proc. SPIE 2552, pp. 556-563 (1995).        R. W. Gooch, T. R. Schimert, W. L. McCardel, and B. A. Ritchey, “Microbolometer and Methode for Forming,” U.S. Pat. No. 6,689,014 (Feb. 10, 2004)        J-J. Yon, A. Astler, M. Vilain, “Thermal Electromagnetic Radiation Detector Comprising an Absorbent Membrane Fixed in Suspension,” U.S. Pat. No. 7,294,836 (Nov. 13, 2007).        G. D. Skidmore, C. G. Howard, “Pixel Structure Having an Umbrella Type Absorber with One or More Recesses or Channels Sized to Increase Radiation Absorption,” U.S. Pat. No. 7,622,717 (Nov. 24, 2009)Phase-Change Microbolometers:        Fred Volkening, “Transition Edge Detector Technology for High Performance IR Focal Plane Arrays,” U.S. Pat. No. 6,576,904 (Jun. 10, 2003).Superconducting Transition-Edge Microbolometers:        K. D. Irwin, and G. C. Hilton, “Transition-Edge Sensors” in Cryogenic Particle Detection edited by C. Enss, Topics Appl. Phys. 66, pp. 63-149 (2005).        K. D. Irwin, “Phonon-Mediated Particle Detection Using Superconducting Tungsten Transition-Edge Sensors,” Thesis Dissertation to Stanford University Department of Physics, 1995.        K. D. Irwin, “An application of electrothermal feedback for high resolution cryogenic particle detection,” Appl. Phys. Lett., 66, pp. 1998-2000 (1995).        M. Galeazzi, and D. McCammon, “Microcalorimeter and bolometer model,” J. Appl. Phys., 93(8) pp. 4856-4869 (2003).        M. J. M. E. de Nivelle, M. P. Bruijn, R. Dde Vries, J. J. Wijnbergen, P. A. J. de Korte, S. Sanchez, M. Elwenspoek, T. Heidenblut, B. Schwierzi, W. Michalke, and E. Steinbeiss, “Low noise high-Tc superconducting bolometers on silicon nitride membranes for far-infrared detection,” J. App. Phys., 82(10) pp 4719-4726 (1997).        I. A. Khrebtov, K. V. Ivanov, and V. G. Malyarov, “Noise properties of high-Tc superconducting transition edge bolometers with electrothermal feedback,” Proc. SPIE v. 6600, paper 660014 (2007).        Irwin et al., “Application of Electrothermal Feedback for High Resolution Cryogenic Particle Detection using a transition edge sensor,” U.S. Pat. No. 5,641,961.        
High sensitivity can be achieved in bolometer or microbolometer by exploiting a “transition edge” effect, whereby some property that changes rapidly over a small temperature range offers a way to measure the property (e.g. resistance) with high sensitivity despite the change in temperature being small. For example, superconducting transition-edge sensors exhibit several orders of magnitude change in conductivity across the transition edge between superconducting and normal resistivity states, and the width of the transition edge can be narrower than 1 kelvin wide. Also, many sources of thermal noise are strongly reduced because some superconducting transition edge sensors operate at very low temperatures (typically below 4.2 K). High performance is still achieved when using high temperature superconductors at temperatures around 77 K and phase change materials at higher temperatures (e.g. near 300 K or room temperature). Transition edge sensors often take advantage of an electrothermal feedback loop that stabilizes the operational temperature of the device and speeds the effective thermal time constant. Additional References describing aspects of transition edge devices and electrothermal feedback include:                M. Galeazzi, “An external electronic feedback system applied to a cryogenic μ-calorimeter,” Rev. Sci. Instrum., 69(5) pp. 2017-2023 (1998).        G. Neto, L. Alberto, L. de Almeida, A. M. N. Lima, C. S. Moreira, H. Neff, I. A. Khrebtov, and V. G. Malyarov, “Figures of merit and optimization of a VO2 microbolometer with strong electrothermal feedback,” Optical engineering 47(7) paper 073603 (2008).        S. H. Moseley, J. C. Mather, and D. McCammon, “Thermal detectors as x-ray spectrometers,” J. Appl. Phys. 56(5) pp. 1257-1262 (1984).        M. Buhler, E. Umlauf, and J. C. Mather, “Noise of a bolometer with vanishing self-heating,” Nuclear Instruments and Methods in Physics Research A 346, pp. 225-229 (1994).        J. C. Mather, “Bolometer noise: nonequilibrium theory,” Applied Optics. 21(6) pp. 1125-1129 (1982).        
Reference is now made to FIG. 2, which shows a diagram prior art microbolometer pixel 24. A microbolometer pixel includes a temperature sensing element 23, an absorber 21, which is designed to absorb an incident electromagnetic signal 22 (denoted PEM) with good efficiency and couple the PEM into 23, and connections 25A and 25B to the substrate. Connections 25A and 25B provide mechanical support to isolate the absorber 21 and temperature sensing element 23 from substrate 29. Connection 25A includes mechanical support member 27A and an electrical connection 28A. Connection 25B includes mechanical support member 27B and electrical connection 28B. Those skilled in the art will recognize that a microbolometer must have, at a minimum, one mechanical support member and two electrical connections; the connections to the substrate illustrated above can be generalized to include more mechanical support and more electrical connections, especially for bridge measurement techniques. Additionally, as illustrated in FIG. 2, electrical connections 28A and 28B will have an associated resistance 33A (denoted R3) and 33B (denoted R4), respectively. Electrical current 32A (denoted I3) flows through resistance 33A, and electrical current 32B (denoted I4) flows through resistance 33B. The voltage of the temperature sensing element 23 is 31A (denoted V3) at connection 25A and 31B (denoted V4) at connection 25B. Connections 25A and 25B provide a thermal link between the temperature sensing element 23 and substrate 29, transferring heat in the form of lattice vibrations (phonons) and energetic electrons (hot electrons). The heat power transferred between the substrate 29 and the sensing element 23 through connection 25A is denoted P1. The heat power transferred between the substrate 29 and the sensing element 23 through connection 25B is denoted P2. In order to achieve good sensitivity to the incident electromagnetic power PEM it is advantageous for the thermal conductivity G to substrate 29 be small. Thermal conductivity G is dependent on P1 and P2, which are influenced by the thermal conductivity of connections 25A and 25B. In addition to thermal conductions through connections 25A and 25B, microbolometer pixel 24 will exchange black-body photons 30 (denoted PBB) with its surroundings, which sets a lower bound on the thermal conductivity.
Absorption of incident electromagnetic energy 22 occurs in absorber 21. A microbolometer may use any absorber 21 that efficiently converts incident electromagnetic energy 22 into thermal energy, since this thermal energy raises the temperature measured at and by sensing element 23. Efficient absorption is assisted by minimizing reflection and by providing a material with a high density of free electrons, such as a thin metallic layer of TiN, NiCr, Ti, Mb, and their oxides. The thickness of the metallic absorber is chosen to have an impedance matching that of free space (approximately 377 Ω), which works out to a typical thickness between 1 nm and 50 nm. Absorption is generally increased by placing the absorber in an optical cavity tuned to achieve high absorption of the wavelengths of interest.
An alternative to using a metallic thin-film absorber is to use an antenna structure to absorb the incident electromagnetic energy 22, converting the electromagnetic energy into a current (see for example: Bluzer et al., U.S. Pat. No. 7,439,508, (Oct. 21, 2008) and S -W Han and D. P. Neikirk, “Design of infrared wavelength-selective microbolometers using planar multimode detectors,” Proc. SPIE 5836, pp. 540-557 (2005)). This current is used to heat a resistive element placed on or near the temperature sensing element 23. Note that in the antenna coupled designs, it is still necessary to maintain high thermal isolation between the temperature sensing element 23 and substrate 29, so the electrical and mechanical connections to the antenna should be designed to minimize thermal conductivity to substrate 29.
State of the art microbolometers for imaging long wave infrared (LWIR) typically use microbolometer pixels 24 with a pixel pitch of 15-50 μm. Each microbolometer pixel 24 has a thermal heat capacity C and a thermal conductivity to its surroundings of G. The incident electromagnetic power PEM causes the temperature of the microbolometer pixel 24 to increase with respect to the case PEM=0). Typically, the temperature of microbolometer pixel 24 will be higher than that of substrate 29, so heat will be transferred to the substrate, with the thermal conductivity described by the parameter G. The time constant, τ for temperature changes of microbolometer pixel 24 is generally estimated from τ=C/G. State of the art microbolometers typically achieve thermal time constants of 1-100 msec, with G between 10 nW/K and 100 nW/K and C between 10 pJ/K and 1000 pJ/K.
Several important noise sources are worth enumerating:    1. Temperature fluctuation noise, caused by the quantum fluctuations in the temperature arising from the flow of heat between the absorber and the substrate through the thermal link with thermal conductivity G. This noise scales as √(kT2G), and can only be reduced by lowering G. Note that thermal conductivity includes thermal conduction through connection 25A and 25B, black body photons 30 which exchange energy with the surroundings, and may include a component due to thermal conduction through gas molecules (which is why most microbolometers are placed within a vacuum).    2. In addition to the background limit, there is often a minimum G that can be used in a microbolometer because the system must maintain a minimum thermal time constant (τ=C/G). Since C is often fixed by processing considerations, it is not practical in the prior art to reduce G indefinitely.    3. In addition to the limit of thermal conductivity through connection 25A and 25B, there is a background limit to the NETD, which is caused by the fluctuation noise in the black-body photons 30 being exchanged between the detector and its surroundings. Photons can be exchanged between the microbolometer and the target, between the microbolometer and the camera body, and between the microbolometer and the substrate. This photon exchange has an associated shot noise that can be calculated as shown in FIG. 3.    4. Readout noise limits arise from a number of causes that impact the signal-to-noise ratio of the microbolometer, and therefore the NETD. These noise sources are well described in the literature, so are only mentioned by name here:            a. Johnson noise        b. 1/f noise        c. Current shot noise            5. Gain noise: while most standard resistive microbolometers do not use gain (i.e. amplification within the detector device itself), alternative approaches may use gain mechanisms. In general, gain is not noiseless, and will have an associated excess noise, which is the additional noise (beyond the shot limit) imposed on the signal by the gain process. As such, the excess noise factor degrades the SNR and therefore the NETD.
Reference is now made to FIG. 3. The performance of a microbolometer is often specified in terms of the noise-equivalent temperature difference (NETD), or identically as the noise-equivalent difference in temperature (NEDT). An example of the bound set by the NETD is calculated for a specific microbolometer. FIG. 3 shows that there is a limit to the minimum achievable NETD for a given value of the thermal conductivity. In general, a microblometer will exhibit a degraded (higher) NETD than the limits shown in FIG. 3 because there will be other sources of noise in the microbolometer system including its packaging and optics.
FIG. 3, which shows the relationship between thermal conductivity G and the lower bound on NETD performance. Axis 99 shows the thermal fluctuation noise limit for NETD, with the logarithmic scale running from 0.1 mK to 1000 mK. Axis 98 is the thermal conductivity G, with the logarithmic scale running from 10−10 W/K to 10−4 W/K. The NETD limit is calculated assuming a 25 μm×25 μm pixel size with 100% fill factor on the absorber, f/1.0 optics with 90% transmission, and a 30 Hz bandwidth. Furthermore, the calculation assumes the microbolometer has a 2τ sR field of view for black body irradiative photons from the surrounding camera body. Curve 90 shows the lower bound on NETD for this microbolometer assuming a target temperature of 300 K, a microbolometer temperature of 300 K, and a camera body temperature of 300 K. Curve 91 shows the lower bound on NETD for this microbolometer assuming a target temperature of 300 K, a substrate temperature (supporting the microbolometer) of 77 K, and a camera body temperature of 300 K. Curve 92 shows the lower bound on NETD for this microbolometer assuming a target temperature of 300 K, a substrate temperature of 77 K, and a camera body temperature of 77 K. This lower bound on NETD performance occurs due to the thermal fluctuation noise limit, which scales as √(kT2G). For thermal conductivity values below about 2×10−9 W/K, a further reduction in G does not improve NETD because the lower bound on NETD becomes dominated by the radiative fluctuation noise of black body photons exchanged between the microbolometer pixel and the camera body, as well as the radiative fluctuation noise of black body photons exchanged between the microbolometer pixel and the scene to be imaged. Cooling of the microbolometer and substrate improves the lower bound on NETD since thermal fluctuation noise scales as √(kT2G). Cooling of the camera body reduces the radiative fluctuation noise of the black body photons exchanged between the camera body and the microbolometer.